Radio Frequency Power Delivery System

ABSTRACT

A system and method are provided for delivering power to a dynamic load. The system includes a power supply providing DC power having a substantially constant power open loop response, a power amplifier for converting the DC power to RF power, a sensor for measuring voltage, current and phase angle between voltage and current vectors associated with the RF power, an electrically controllable impedance matching system to modify the impedance of the power amplifier to at least a substantially matched impedance of a dynamic load, and a controller for controlling the electrically controllable impedance matching system. The system further includes a sensor calibration measuring module for determining power delivered by the power amplifier, an electronic matching system calibration module for determining power delivered to a dynamic load, and a power dissipation module for calculating power dissipated in the electrically controllable impedance matching system.

RELATED APPLICATION

This application claims the benefit of U.S. Provisional Application No.60/731,797, filed on Oct. 31, 2005, the entire teachings of which areincorporated herein by reference.

BACKGROUND

Various approaches exist for providing RF power to dynamic loads. RFgenerators provide power to dynamic loads typically at frequenciesbetween about 400 kHz and about 200 MHz. Frequencies used in somescientific, industrial and medical applications are approximately 2 MHz,13.56 MHz and 27 MHz.

As shown in FIG. 1A, one system 100 for providing RF power to dynamicloads (i.e., a plasma load 140) involves a fixed frequency RF generator110 and a two-axis tunable matching network 120 connected by a 50 Ωtransmission line 130. The tunable matching network 120 includes aseries motorized vacuum variable capacitor 122 and inductor 124 and ashunt motorized vacuum variable capacitor 126. The algorithm used todetermine the series and shunt capacitance is based on impedancemeasurements typically made using a magnitude and phase detector 150.Independent power control is based on power measurements at the RFgenerator 110. The power control loop 160 and impedance control loop 162are independent.

As shown in FIG. 1B, another system 100′ for providing RF power todynamic loads involves a fixed element matching network 120′ fed by anRF generator 110 and connected by a 50 Ω transmission line 130. Thefixed element matching network 120′ includes a series capacitor 122 andinductor 124 and a shunt capacitor 126. The frequency of the RFgenerator 110 can be tuned to a certain range (e.g., 13.56 MHz ±5%). TheRF generator 110 frequency command is based on the value of voltagestanding wave ratio (VSWR). The independent power loop and VSWR(impedance) control loop 160′ are based on measurements at the output ofthe RF generator 110.

As shown in FIG. 1C, another system 100″ for providing RF power todynamic loads involves an integrated RF generator-impedance matchingnetwork 120″. The RF generator-impedance matching network 120″ includesa series capacitor 122 and inductor 124 and a plurality of shuntcapacitor 126 a . . . 126 n. The shunt capacitor 126 a . . . 126 n arecoupled to a switching circuit 127 a . . . 127 n that couples anddecouples the capacitors 126 to ground. The power control and frequencycontrol 160″ of the system 100″ are not conducted simultaneously.

SUMMARY

These prior art techniques and methods have disadvantages. Higher costis typically associated with prior art techniques and methods due to theneed for at least two separate modules: 1) the RF generator/amplifierand 2) the impedance matching network, which are to be connected via atransmission line. Furthermore, each module requires a RFvoltage/current sensor or a magnitude/phase detector.

Plasma impedance is a function of the power delivered to the plasma.Furthermore, the power delivered by the RF generator is a function ofthe impedance “seen” by the generator. As a result, a clear circularinterdependence exists between delivered power and load impedanceyielding a multi-input-multi-output (MIMO) system with cross-coupling.In prior art systems, the RF generator control loop and the impedancematching control loop are independent and thus cannot compensate for thecross-coupling between power control and impedance matching controlloops. This leads to poor closed-loop performance.

The dynamic response of any controlled system is only as fast as theslowest functional module (sensor, actuator, or control systemparameters). In prior art systems, the slowest functional module istypically the DC power supply. Specifically, the DC power supplied tothe input of the RF power amplifier usually includes a largeelectrolytic capacitor that is used to filter higher frequencies. Thedownside of using such a filter network is that the dynamic response(e.g., response to a step change in power command) is slow regardless ofthe control update rate. The system is therefore unable to sufficientlycompensate for plasma instabilities.

In systems that use a vacuum capacitor driven by motors, the responsetime is on the order of hundreds of milliseconds. Owing to the fact thatplasma transients (sudden and rapid change of impedance) of interestoccur within hundreds of microseconds, the vacuum capacitor cannot beused to match load changes attributed to plasma transients.

Control algorithms for matching networks used in the prior art haverelied upon the real and imaginary components of the measured impedance.Impedance measurement-based matching control suffers from an inherentdisadvantage. For example, a change in shunt capacitance to correct ormodify the real component of the impedance results in an undesirablechange in the imaginary component of the impedance. Similarly, a changein the series capacitance or frequency to correct or modify theimaginary component of the impedance results in an undesirable change inthe real component of the impedance. The matrix that relates thecontrolled variable vector (formulated by the real and imaginarycomponents of the impedance) and the controlling variable vector(formulated by the shunt and series capacitance or the shunt capacitanceand frequency) is non-diagonal. Impedance measurement-based controlalgorithms are therefore not effective. Control algorithms based on theimpedance formulated by using magnitude and phase measurements of theimpedance are similarly ineffective.

Calibration methods for prior art systems calibrate the RF impedanceanalyzer or VI probe at the input of the electronic matching network.These calibration methods assume the power loss in the electronicmatching network is fixed for all states of the electronic matchingnetwork and operating frequencies. However, the losses of the electronicmatching network contribute significantly to the overall systemoperation.

Accordingly, a need therefore exists for improved methods and systemsfor controlling power supplied to a dynamic plasma load and the lossesassociated therewith.

There is provided a system for delivering power to a dynamic load. Thesystem includes a power supply providing DC power having a substantiallyconstant power open loop response, a power amplifier for converting theDC power to RF power, a sensor for measuring voltage, current and phaseangle between voltage and current vectors associated with the RF power,an electrically controllable impedance matching system to modify theimpedance of the power amplifier to at lease substantially match animpedance of a dynamic load, and a controller for controlling theelectrically controllable impedance matching system. The system furtherincludes a sensor calibration measuring module for determining powerdelivered by the power amplifier, an electronic matching systemcalibration module for determining power delivered to a dynamic load,and a power dissipation module for calculating power dissipated in theelectrically controllable impedance matching system.

In one embodiment, the electrically controllable impedance matchingsystem can include an inductor, a capacitor in series with the inductor,and a plurality of switched capacitors in parallel with the dynamicload. The inductor can be a multiple tap-type inductor or avariable-type inductor. Each of the plurality of switched capacitors canbe in series with a switch and an additional capacitor. In anotherembodiment, the electrically controllable impedance matching system caninclude a capacitor, and a plurality of switched capacitors in parallelwith the dynamic load, wherein each of the plurality of capacitors is inseries with a switch and an additional capacitor. In yet anotherembodiment, the electrically controllable impedance matching system cancontrol the frequency of the impedance matching between the poweramplifier and the dynamic load.

In one embodiment, the controller can control the electricallycontrollable impedance matching system for simultaneous control ofconductance and susceptance associated with the impedance between thepower amplifier and the dynamic load. In another embodiment, thecontroller can simultaneously control RF power frequency, RF powermagnitude and the impedance between the power amplifier and the dynamicload. In yet another embodiment, the controller can control theelectrically controllable impedance matching system for regulatingconductance and susceptance to setpoints that stabilize an unstabledynamic load.

The power dissipated in the electrically controllable impedance matchingsystem is the difference between the power delivered by the poweramplifier and the power delivered to the dynamic load. The powerdelivered to the dynamic load is a sum of the power delivered to aresistive load and the power dissipated inside the load simulator.

The sensor calibration measuring module calibrates the sensor into aresistive load, wherein the resistive load is 50 Ω. The electronicmatching module calibrates an output of the electrically controllableimpedance matching system into a load simulator. The load simulator canbe an inverse electrically controllable impedance matching system. Theelectronic matching system calibration module can include a power metercalibration module for determining power delivered to a resistive load;and a load simulator calibration module for determining power dissipatedinside the load simulator. The resistive load can be 50 Ω. The radiofrequency power delivery system provides at least the followingadvantages over prior art systems. The system can enhance power setpointregulation, impedance matching, and load disturbance mitigation usinghigh-speed (e.g., in excess of 50 kHz in one embodiment) digitalmulti-input-multi-output (MIMO) control. The system can operate in thepresence of transient changes in plasma load properties and underconditions involving fast plasma stabilization. The system can provide aRF power delivery system that is robust to transients during startup ofthe system. The system can provide a high power step-up ratio, whereinthe high power step-up ratio is 100 (e.g., 15 W to 1500 W). The systemcan measure power delivered to the load connected to the output of theintegrated generator system. The system can allow for regulation ofpower that is independent of the power loss variation associated withthe state/value of various controlled variables. The system caneliminate the need for recipe-based calibration for plasma loads.

BRIEF DESCRIPTIONS OF THE DRAWINGS

The foregoing and other objects, features and advantages of theinvention will be apparent from the following more particulardescription of preferred embodiments of the invention, as illustrated inthe accompanying drawings. The drawings are not necessarily to scale,emphasis instead being placed upon illustrating the principles of theinvention.

FIG. 1A is a diagram of an RF power delivery system having a two-axistunable matching network according to the prior art;

FIG. 1B is a diagram of an RF power delivery system having a fixedmatching network according to the prior art;

FIG. 1C is a diagram of an RF power delivery system having an integratedRF generator-impedance matching network according to the prior art;

FIG. 2 is a module-based diagram of the On-Chamber RF power deliverysystem;

FIG. 3 is a plasma stability graph;

FIG. 4 is one embodiment of a fast DC bus of FIG. 2;

FIG. 5 is one embodiment of an RF impedance analyzer or VI Probe of FIG.2

FIG. 6 is one embodiment of an electronic matching network of FIG. 2;

FIG. 7 is one embodiment of a module-based diagram of a DSP compensatorboard of FIG. 2;

FIG. 8 is a block diagram for calibrating the On-Chamber RF powerdelivery system;

FIG. 9A is one embodiment for calibrating a power meter to a 50 Ωcalorimeter power reference;

FIG. 9B is one embodiment for calibrating a load simulator to a DC powerreference;

FIG. 9C is one embodiment for calibrating an RF impedance analyzer intoa 50 Ω load; and

FIG. 9D is one embodiment for calibrating power delivered into the loadsimulator.

DETAILED DESCRIPTION

Generally, an integrated radio frequency (RF) power delivery system isprovided for dynamic load applications (e.g., inductive and/orcapacitive plasma load). FIG. 2 is an illustration of the integratedradio frequency (RF) power delivery system 200. Representativefunctional modules of the integrated system 200 include a fast DC bus210, an RF power amplifier (“PA”) 220, a digital signal processor(“DSP”) compensator board 230, an RF impedance analyzer or VI probe 240,and an electronic matching network 250. The system 200 is coupled to aplasma load 260. It should be understood by one skilled in the art thatthe integrated system 200 can be implemented for a wide range ofresistive and reactive loads.

Generally, the fast DC bus 210 delivers DC power to the power amplifier220. The power amplifier 220 converts the DC power from the fast DC bus210 to an RF power at a frequency. The electronic matching system 250switches shunt capacitors (not shown) to match the impedance between thepower amplifier 220 and the plasma load 260 to facilitate stable andmaximum power transfer from the power amplifier 220 to the plasma load260. The DSP compensator board 230 controls the operation of the system200 based on measurements received from the fast bus controller 212 andRF impedance analyzer 240. The RF impedance analyzer 240 measures theRMS voltage, RMS current, and phase angle between the RF voltage andcurrent vectors. Based on these measurements, relevant RF parameters arecomputed by the DSP compensator board 230. These parameters include, butare not limited to impedance vector z, admittance vector y, deliveredpower P_(del), and voltage-standing wave ratio (“VSWR”). Typicaloperations of the DSP compensator board include power setpoints throughthe fast bus controller 212, RF power frequency setpoints through thepower amplifier driver 222, and switching frequency through theelectronic match controller 252.

In one aspect, the system 200 achieves simultaneous power and impedanceregulation. Independent susceptance regulation allows for theimplementation of a frequency control algorithm based only on thedeviation of the conductance from the conductance setpoint. As a result,both control loops can be operated simultaneously and at high-speedresulting in improved robustness. Further, well-known instabilities forelectronegative plasmas at low-pressure (e.g., SF6 at 5 mT at 300 W asillustrated in FIG. 3) can be stabilized by setting arbitraryconductance and susceptance setpoints in conjunction with operation ofthe Fast DC bus 210.

FIG. 4 is a diagram of a partial resonant inverter power supply typefast DC bus 210. The fast DC bus 210 provides process stability due toits associated constant power open loop response. The fast DC bus 210improves FET utilization over the entire load space which results inmore power being delivered to the load with the same PA 220 (FIG. 2).The fast DC bus 210 has a fast response rate allowing it to deliverincreased power to the plasma so it does not extinguish while alsoallowing the flexibility to reduce the bus voltage to ensure the FETs onthe PA 220 operate in a safe mode. Other types of topologies can for thefast DC bus 210 can be used. See for example, co-pendingcontinuation-in-part application it's parent U.S. application Ser. No.10/947,397 filed Sep. 22, 2004, the entire teaching of each applicationare herein incorporated by reference.

In one embodiment, the fast DC bus can be a partial resonant inverter210 that includes a pair of switches (MOSFETs) 302 a, 302 b, an inductor(L) 306, a capacitor (C) 308, and four diodes 310 a, 310 b, 310 c, and310 d. In operation, the partial resonant inverter 300 converts theinput voltage into a square wave or other known type DC wave form. Thesquare wave is passed through the inductor 306 and capacitor 308, thecombination of which form an LC filter, clamped by the diodes 310 c, 310d, coupled and rectified by a transformer rectifier 304 and filtered toobtain a desired DC voltage (power setpoint). The DC power setpoint isprovided from the DSP compensator board 230 (FIG. 2). The desiredimpedance setpoint can be specified in terms of its vector inverse(referred to as admittance) and which constitutes simultaneousregulation of conductance to an arbitrary conductance setpoint andregulation of susceptance to an arbitrary susceptance setpoint. Theoutput of the partial resonant inverter 300 (DC-DC converter) isconnected to DC input of the RF power generator/amplifier 220.

In operation, the capacitor 308 is periodically charged to an input railvoltage (+Vin) and discharged while the capacitor current is passed viathe plasma load 260 (FIG. 2). Every charge or discharge cycle, theenergy deposited in the resistive load is equal to CV²/2, independent ofload resistance. Thus, the power is equal to F_(SW)×CV²/2, where F_(SW)is the switching frequency and V is the input voltage. The inductor 306ensures that the capacitor 308 is fully charged and discharged in finitetime. One advantage of the partial resonant inverter 300 design is theability to control the output voltage by varying either V or/and F_(SW).

FIG. 5 is a diagram of one embodiment of an RF impedance analyzer or VIProbe 240. The VI Probe 240 includes a DC power supply 242, an analysisboard assembly 244, and a probe head assembly 246. The analysis boardassembly 244 receives low-level RF signals from the probe head assembly246. The probe head assembly 246 provides two voltage outputs: 1) avoltage representation of the time varying electric field present in theprobe head assembly 246 (voltage signal); and 2) a voltagerepresentation of the time varying magnetic field present in the probehead assembly 246 (current signal). The analysis board assembly 244receives and processes the two voltage outputs of the probe headassembly 246 and outputs the RF parameters to the DSP compensator board230 (FIG. 2). MKS Instruments, Inc. VI-Probe-4100 and VI-Probe-350 areexemplary analyzers that can be used for this purpose.

FIG. 6 is a diagram of one embodiment of an electronic matching network250. In one embodiment, the electronic matching 250 includes aninductance 254 in series with the load 260 (e.g., a compact inductorwith multiple tap points), a fixed or variable series-padding capacitor252, and field effect transistors (“FET's”) 256 a . . . 256 n thatswitch one or more upper capacitors C_(tu)(i) 258 a . . . 258 n to acorresponding lower capacitor C_(td)(i) 258 a′ . . . 258 n′, which isterminated to ground. In some embodiments, the electronic matching 250network does not include the inductance 254 in series with the load 260.Other types of electronic matching networks can be used. See forexample, U.S. Pat. No. 6,887,339, the entire teaching of which is hereinincorporated by reference.

FIG. 7 shows a module-based diagram of a DSP compensator board 230. TheDSP compensator board 230 incorporates both a digital signal processor(“DSP”) and a field programmable gate array (“FPGA”), and togethercontrols the entire integrated system 200. The DSP compensator boardincludes an admittance compensation module 232, a frequency controlmodule 234, an electronic match control module 236, an RF powercomputation module 237, and an RF power control module 238. Generally,the DSP compensator board receives the output from the VP probe 240. Theadmittance computation module 232 uses the VI probe outputs to calculatethe admittance of the system 200. The frequency control module 234 usesthe admittance to vary the frequency of the power amplifier 220. Theelectronic match control module 236 uses the admittance to switch theFETs 256 of the electronic matching network 250 on or off. The RF powercomputation module 237 uses the VI probe outputs to calculate the RFpower of the system 200. The RF power control module 234 uses the RFpower computation to regulate the power supplied from the fast DC buspower 210. A more detailed description of the operation of the system200 is set forth below.

One embodiment of the power regulation objective and algorithm is setforth below: The objective is to regulate the delivered power P_(del) toa user-defined setpoint P_(sp). To ensure smooth transitions, trajectorygenerators are used. In one embodiment, a first-order trajectory isgenerated as follows: $\begin{matrix}{\frac{\mathbb{d}P_{t}}{\mathbb{d}t} = {\frac{1}{\tau_{t}}( {{P_{t}(t)} - P_{sp}} )}} & {{EQN}.\quad 1}\end{matrix}$where τ_(t) is the trajectory time constant and P_(t) is the desiredpower trajectory. The delivered-power control algorithm, in terms of thechange in power commanded to the Fast Bus, is given by the followingrelationship:P _(cmd) =k _(p)(P _(t) −P _(del))+k _(i)∫(P _(t) −P _(del))dt  EQN. 2where k_(P) and k_(i) are the proportional and integral gains,respectively.

Admittance regulation objective: A normalized admittance vector isdefined as follows: y=g+jb where g is the normalized conductance and bis the normalized susceptance. The impedance matching control objectiveis formulated as follows: g→g_(sp) and b→b_(sp) where g_(sp) and b_(sp)are arbitrary setpoints selected to improve plasma stability. The aboveobjective is reinterpreted in terms of impedance by noting thatimpedance is defined as the reciprocal of admittance, according to thefollowing relationship: $\begin{matrix}{z = {\frac{1}{y} = {{r + {j\quad x}} = {\frac{R + {j\quad X}}{Z_{0}} = \frac{R + {j\quad X}}{R_{0} + {j\quad 0}}}}}} & {{EQN}.\quad 3}\end{matrix}$where z is the normalized impedance, r and x are the resistance andreactance, respectively, Z₀=R₀+j0 denotes a nominal RF amplifiercharacteristic impedance. It follows that when g→1 and b→0, we obtainR→R₀ and X→0.

Admittance regulation algorithm: The frequency control loop is designedby using conductance measurements, for example, as a PI controlalgorithm as follows:f _(tcmd) =−k _(pf)(g _(sp) −g)−k _(if)∫(g _(sp) −g)dt  EQN. 4where k_(pf) and k_(if) are scalar proportional and integral controlgains. The shunt capacitance control loop is designed by usingconductance measurements, for example, as a PI control algorithm asfollows:C _(tcmd) =−k _(pc)(b _(sp) −b)−k _(ic)∫(b _(sp) −b)dt  EQN. 5where d_(pc) and k_(ic) are scalar proportional and integral controlgains.

In operation, referring now to FIGS. 2, 3 and 6, after the user providesa non-zero setpoint, the trajectory generator and the power andadmittance control algorithms are simultaneously activated and executed.The VI probe 240 provides analog signals proportional to the RF voltageand RF current, which are synchronously sampled by the analog-to-digitalconverters, sent to a mixer and CIC filter (not shown) and ultimatelysent through a calibration matrix to yield RF voltage and RF currentmeasurements given by the following relationships:V =V_(r) +jV _(i)andI =I_(r) +jI _(i)  EQN. 6where V, I denote vector representations of the instantaneous RF voltageand current, respectively, and subscripts r and i are used to denote thescalar values of the real and imaginary components.

The average delivered power is computed as follows: $\begin{matrix}{P_{del} = {{\frac{1}{2}{Re}\{ {\overset{\_}{VI}}^{*} \}} = {{V_{r}I_{r}} + {V_{i}I_{i}}}}} & {{EQN}.\quad 7}\end{matrix}$where Re{} denotes the real component of the vector, and superscript *is used to denote the complex conjugate of the vector.

The admittance vector Y is then computed as follows: $\begin{matrix}{\overset{\_}{Y} = {\frac{\overset{\_}{I}}{\overset{\_}{V}} = {{\frac{( {{I_{r}V_{r}} + {I_{i}V_{i}}} )}{V_{r}^{2} + V_{i}^{2}} + {j\frac{( {{I_{i}V_{r}} - {I_{r}V_{i}}} )}{V_{r}^{2} + V_{i}^{2}}}} \equiv {G + {j\quad B}}}}} & {{EQN}.\quad 8}\end{matrix}$where the conductance G and the susceptance B are real and imaginarycomponents of the admittance Y.

The normalized conductance g and nonnalized susceptance b are computedas follows: $\begin{matrix}{{g = {{Z_{0}G} = {Z_{0}\frac{( {{I_{r}V_{r}} + {I_{i}V_{i}}} )}{V_{r}^{2} + V_{i}^{2}}}}}\quad{and}\quad{b = {{Z_{0}B} = {Z_{0}\frac{( {{I_{i}V_{r}} - {I_{r}V_{i}}} )}{V_{r}^{2} + V_{i}^{2}}}}}} & {{EQN}.\quad 9}\end{matrix}$where Z₀ denotes the characteristic impedance of the RF amplifier. Themeasurements of P_(del), g, b are respectively sent to the controlalgorithms for P_(cmd), f_(cmd), C_(tcmd) respectively.

The electronic match controller 252 switches the FETs 256 (FIG. 6)thereby switching the shunt capacitors 258 to match the impedancebetween the power amplifier 20 and the dynamic load 260. The absence ofmoving mechanical parts leads to higher reliability. In one embodiment,the step response of the system 200 is faster than about 1 ms becausethe speed of the response is governed by the electronics and not by themechanical response.

A change in frequency results in a change in both the conductance andthe susceptance. However, for an integrated system without transmissionline cables, a change in shunt capacitance results only in a change inthe susceptance and does not affect the conductance value. Thus, thematrix that relates the controlled variable vector (formulated by thereal and imaginary components of the admittance) and the controllingvariable vector (formulated by the shunt and series capacitance or theshunt and frequency) is triangular. As a result, independent susceptanceregulation is achieved by varying the shunt capacitance.

Independent susceptance regulation allows for the implementation of afrequency control algorithm based only on the deviation of theconductance from the conductance setpoint. As a result, both theconductance-based frequency control loop and the susceptance-based shuntcapacitance control loop can be operated simultaneously and athigh-speed, resulting in improved robustness.

FIG. 8 is a block diagram 300 of a method for determining the powerdissipated (loss) in the electronic matching network 250 (FIG. 2) toimprove the efficiency of the system 200. Step one (310), a power meter314 (FIG. 9A) is calibrated into a 50 Ω calorimeter power reference todetermine the power delivered to the 50 Ω load. Step two (320), a loadsimulator calorimeter 332 (FIG. 9B) is calibrated to a DC powerreference to determine the power dissipated inside a load simulator 342(FIG. 9D). Step three (330), the VI probe 240 (FIG. 2) is calibratedinto a 50 Ω load to determine the power delivered by the power amplifier220 (FIG. 2). Step four (340), the output of the system 200 iscalibrated into the load simulator 342 to determine the power deliveredto Z_(L)=R_(L)+jX_(L). Step 5 (350), the power dissipated in theelectronic matching system is calculated by difference between the hepower delivered by the power amplifier 220 and the power delivered to.Z_(L)=R_(L)+jX_(L).

FIG. 9A is detailed implementation diagram of step 310 for calibratingthe power meter 314. A calorimeter 322 is coupled to the output of theVI Probe 240, RF power is applied from the power amplifier 220, and thepower meter 314 is calibrated. Calorimetry is the measurement of thermallosses. It is implemented by thermally insulating the 50 Ω load in thecalorimeter (322) to prevent ambient thermal losses and measuring theflow rate and the temperature rise of the cooling water. The power meteris calibrated to the power dissipation in the load computed by${Q = {\frac{\mathbb{d}m}{\mathbb{d}t}{C( {T_{out} - T_{i\quad n}} )}}},\quad{{where}\quad\frac{\mathbb{d}m}{\mathbb{d}t}}$denotes the mass flow rate, C denotes the specific heat of water, andT_(in), T_(out) denote the inlet and outlet temperatures, respectively.A computer 324 acquires flow rate and temperature measurements tocompute the power dissipation in the load and the difference (error)with respect to readout of the power meter. The computer 324 thenapplies this error as a correction to the power meter to complete thecalibration.

FIG. 9B is detailed implementation diagram of step 320 for calibratingthe load simulator calorimeter 332. A load simulator calorimeter 332 iscoupled to a DC power supply 334, DC power is applied, and the loadsimulator calorimeter 332 is calibrated. The DC power supply providesthe DC power measurements. Using flow rate and temperature measurementsat the inlet and outlet of the cooling system, a computer 324 computesthe power dissipated in the load simulator. The computer 324 thenapplies the error between the power reported by the DC power supply andthe power computed using calorimetry as a correction to the loadsimulator to complete the calibration.

FIG. 9C is detailed implementation diagram step 330 for calibrating anRF impedance analyzer or VI probe 240. Generally, the VI Probe 240calibration in each integrated RF generator system 200 includes thefollowing steps that yield a matrix transfer function that relates theVI probe voltage and current measured by the DSP compensator board 230to an actual RF line voltage and current.

First, a short circuit connector 312 is coupled to the RF line outputterminal of the VI probe 240, RF power is applied from the poweramplifier 220, and Z^(dsp) _(sc) is computed, wherein Z^(dsp) _(sc) isdefined as the ratio of V_(dsp)/I_(dsp) as measured by the DSPcompensator board 230 for short circuit. Second, an open circuitconnector 314 is coupled to the RF line output terminal of the VI probe240, RF power is applied from the power amplifier 220, and Z^(dsp) _(ac)is computed, wherein Z^(dsp) _(ac) is defined as the ratio ofV_(dsp)/I_(dsp) as measured by the DSP compensator board 230 for opencircuit. Third, a 50 Ω load (Z_(L)) 316 is coupled to the output of theVI Probe 240, RF power is applied from the power amplifier 220, V_(m)and I_(m) are recorded and the RF line voltage V_(L) is computed,wherein V_(L)=√{square root over (P_(L)Z_(L))}·P_(L) is the deliveredpower measured by a power meter 318 at the 50 Ω load 316. Lastly, the VIprobe calibration matrix transfer function is computed by the followingequation: $\begin{matrix}{\lbrack \frac{V_{L}(t)}{I_{L}(t)} \rbrack = {\begin{pmatrix}\frac{V_{L}}{V_{m} - {Z_{sc}^{dsp}I_{m}}} & \frac{{- Z_{sc}^{dsp}}V_{L}}{V_{m} - {Z_{sc}^{dsp}I_{m}}} \\\frac{- V_{L}}{Z_{L}( {{{- Z_{sc}^{dsp}}I_{m}} - V_{m}} )} & \frac{Z_{sc}^{dsp}V_{L}}{Z_{L}( {{{- Z_{sc}^{dsp}}I_{m}} - V_{m}} )}\end{pmatrix}\lbrack \frac{V_{dsp}(t)}{I_{dsp}(t)} \rbrack}} & {{EQN}.\quad 10}\end{matrix}$

The expression in equation 10 translates VI probe measurement signalsinto RF line voltage and RF line current at the output of the VI probe240.

FIG. 9D is detailed implementation diagram step 340 for calibrating thesystem 200 (FIG. 2). The system level calibration is used to quantifythe power loss in the electronic matching network 250 for a range ofvalues matching network variables. A load simulator 342 is coupled tothe output of the electronic matching network 250. Typically, the loadsimulator is an electronic matching network inverse to the electronicmatching network 250. A 50 Ω load is coupled to the output of the loadsimulator 342. The system-level calibration of the RF generator system200 is performed as follows. First, a series inductance is adjusted inll steps for L_(S)ε[L_(s min), L_(s max)]. Second, a power setpointvalue is changed in pp steps P_(sp)ε[P_(sp min), P_(sp max)] W. Third, ashunt capacitance setpoint value is changed in cc stepsC_(tcmd)ε[C_(tcmd min), C_(tcmd max)]. Lastly, an RF frequency value ischanged in ff steps fε[f_(min), f_(max)] Hz.

For each combination of the aforementioned steps, the load simulator 342is set to present an impedance mismatch at the output of the electronicmatching network 250. Next, RF power is applied from the power amplifier220 and the power meter 314 measures the terminating load 312resistance. The terminating load resistance is denoted by P_(50Ω) andtransformed to the input of the load simulator 342. The simulated loadis denoted by P_(sys) as P_(sys)=f_(50-to-sim)(P_(50Ω), C₁, C₂), whereC₁, and C₂, represent the series and shunt capacitance of the loadsimulator and f_(50-to-sim) represents a tabular arrangement. The lossesassociated in electronic matching network 250 is computed by thedifference between the P_(L) and P_(50Ω).

In some embodiments, a calibration table which has dimensionsll×pp×cc×ff can stored in non-volatile memory (e.g., flash memory) asP_(sys)=f_(VI-to-sim)(L_(s), P_(sp), C_(tcmd), f), where f_(VI-to-sin)represents a tabular arrangement. High-speed real-time control loopsnecessitate fast searches through the calibration table during operationof the system 200. Non-volatile memory (e.g., flash memory) tends to beslower than the volatile memory (e.g., Dynamic RAM). The high-speedvolatile memory is effectively utilized, wherein the arrangement of thecalibration table (dimensions ll×pp×cc×ff) can be based on howfrequently L_(s), P_(sp), C_(tmcd), and f are changed. Specifically, thecalibration table can be segmented into ll memory blocks; each blockincluding pp memory pages; each memory page including a cc×ffdimensional table. A new memory block can be loaded into non-volatilememory when L_(s) is changed, a new memory page can be loaded when powersetpoint is changed, and calibration points for the appropriate memorypage associated with C_(tcmd) and f can be executed in real-time.

While this invention has been particularly shown and described withreferences to preferred embodiments thereof, it will be understood bythose skilled in the art that various changes in form and details may bemade therein without departing from the scope of the inventionencompassed by the appended claims.

1. A system for delivering power to a dynamic load, comprising: a powersupply providing DC power having a substantially constant power openloop response; a power amplifier for converting the DC power to RFpower; a sensor for measuring voltage, current and phase angle betweenvoltage and current vectors associated with the RF power; anelectrically controllable impedance matching system to modify theimpedance of the power amplifier to at least substantially match animpedance of a dynamic load; and a controller for controlling theelectrically controllable impedance matching system.
 2. The system ofclaim 1, wherein the controller controls the electrically controllableimpedance matching system for simultaneous control of conductance andsusceptance associated with the impedance between the power amplifierand the dynamic load.
 3. The system of claim 1, wherein the electricallycontrollable impedance matching system comprises an inductor, acapacitor in series with the inductor, and a plurality of switchedcapacitors in parallel with the dynamic load.
 4. The system of claim 3,wherein the inductor is a multiple tap-type inductor or a variable-typeinductor.
 5. The system of claim 3, wherein each of the plurality ofswitched capacitors is in series with a switch and an additionalcapacitor.
 6. The system of claim 1, wherein the electricallycontrollable impedance matching system comprises a capacitor, and aplurality of switched capacitors in parallel with the dynamic load,wherein each of the plurality of capacitors is in series with a switchand an additional capacitor.
 7. The system of claim 1, wherein thecontroller simultaneously controls RF power frequency, RF powermagnitude and the impedance between the power amplifier and the dynamicload.
 8. The system of claim 1, wherein the electrically controllableimpedance matching system controls the frequency of the impedancematching between the power amplifier and the dynamic load.
 9. The systemof claim 1, wherein the controller controls the electricallycontrollable impedance matching system for regulating conductance andsusceptance to setpoints that stabilize an unstable dynamic load. 10.The system of claim 1, further comprising: a sensor calibrationmeasuring module for determining power delivered by the power amplifier;an electronic matching system calibration module for determining powerdelivered to a dynamic load; and a power dissipation module forcalculating power dissipated in the electrically controllable impedancematching system.
 11. The system of claim 10, wherein power dissipated inthe electrically controllable impedance matching system is thedifference between the power delivered by the power amplifier and thepower delivered to the dynamic load.
 12. The system of claim 10, whereinsensor calibration measuring module calibrates the sensor into aresistive load.
 13. The system of claim 12, wherein the resistive loadis 50 Ω.
 14. The system of claim 10, wherein electronic matching systemcalibrates an output of the electrically controllable impedance matchingsystem into a load simulator.
 15. The system of claim 14, wherein theload simulator is an inverse electrically controllable impedancematching system.
 16. The system of claim 10, wherein the electronicmatching system calibration module includes: a power meter calibrationmodule for determining power delivered to a resistive load; and a loadsimulator calibration module for determining power dissipated inside theload simulator.
 17. The system of claim 15, wherein the resistive loadis 50 Ω.
 18. The system of claim 15, wherein the power delivered to thedynamic load is a sum of the power delivered to a resistive load and thepower dissipated inside the load simulator.
 19. A method for deliveringpower to a dynamic load, comprising: providing DC power having asubstantially constant power open loop response; converting the DC powerto RF power through a power amplifier; measuring voltage, current andphase angle between voltage and current vectors associated with the RFpower through a sensor; and modifying the impedance of the poweramplifier to at least substantially match an impedance of a dynamic loadwith an electrically controllable impedance matching system.
 20. Themethod of claim 19, further comprising simultaneously controllingconductance and susceptance associated with the impedance between thepower amplifier and the dynamic load.
 21. The method of claim 19,further comprising simultaneously controlling RF power frequency, RFpower magnitude and the impedance between the power amplifier and thedynamic load.
 22. The method of claim 19, further comprising controllingthe frequency of the impedance matching between the power amplifier andthe dynamic load.
 23. The method of claim 19, further comprisingcontrolling the electrically controllable impedance matching system forregulating conductance and susceptance to setpoints that stabilize anunstable dynamic load.
 24. The method of claim 19, further comprising:determining RF power delivered by the power amplifier; determining powerdelivered to a dynamic load; and calculating power dissipated in theelectrically controllable impedance matching system.
 25. The method ofclaim 24, wherein power dissipated in the electrically controllableimpedance matching system is the difference between the power deliveredby the power amplifier and the power delivered to the dynamic load. 26.The method of claim 24, wherein determining power delivered to a dynamicload includes: determining power delivered to a resistive load; anddetermining power dissipated inside a load simulator.
 27. The method ofclaim 26, wherein the power delivered to the dynamic load is a sum ofthe power delivered to a resistive load and the power dissipated insidethe load simulator.
 28. The method of claim 19, further comprisingcalibrating the sensor into a resistive load.
 29. The method of claim19, further comprising calibrating an output of the electricallycontrollable impedance matching system into a load simulator.
 30. Amethod for delivering power to a dynamic load, comprising: means forproviding DC power having a substantially constant power open loopresponse; means for converting the DC power to RF power through a poweramplifier; means for measuring voltage, current and phase angle betweenvoltage and current vectors associated with the RF power through asensor; and means for modifying the impedance of the power amplifier toat least substantially match an impedance of a dynamic load with anelectrically controllable impedance matching system.
 31. The method ofclaim 30, further comprising: means for determining RF power deliveredby the power amplifier; means for determining power delivered to adynamic load; and means for calculating power dissipated in theelectrically controllable impedance matching system.